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mic24054 12v, 9a high - efficiency buck regulator superswitcher ii ? hyper light load is a registered trademark of micr el, inc. hyper speed control, superswitcher ii, and any capacitor are trademarks of micrel, inc. micrel inc. ? 2180 fortune drive ? san jose, ca 95131 ? usa ? tel +1 ( 408 ) 944 - 0800 ? fax + 1 (408) 474 - 1000 ? http://ww w.micrel.com october 2012 m9999 -102 5 12 -a general description the micrel mic24054 is a con stant - frequency, synchronous dc/ dc buck regulator featuring adaptive on- time control architecture. the mic24054 operates over a supply range of 4.5v to 19v . it has an internal linear regulator which provides a regulated 5 v to power the internal control circuitry. the mic24054 operates at a constant 600khz switching frequency in continuous conduction mode and can be used to provide up to 9 a of output current. the output voltage is adjustable down to 0.8v. micrel?s hyper light load ? architecture provides the same high - efficiency and ultra - fast transient response as the hyper speed control ? architecture under medium to heavy loads, but also maintains high efficiency under light load conditions by transitioning to vari able frequency, discontinuous mode operation. the mic24054 offers a full suite of protection features to ensure protection of the ic during fault conditions. these include undervoltage lockout to ensure proper operation under power - sag conditions, thermal shutdown, internal soft - start to reduce the inrush current, fold back curr ent limit and ?hiccup mode ? short - circuit protection. the mic24054 includes a power good (pg) output to allow simple sequencing. the 9a hyper speed control part, mic24053 , is also av ailable on micrel?s web site. all support documentation can be found on micrel?s web site at : www.micrel.com . features ? hyper light load e fficiency ? up to 80% at 10ma ? hyper speed contro l architecture enables ? hi gh delta v operation (v in = 19v and v out = 0.8v) ? small output capacitance ? inp ut voltage range: 4.5v to 19v ? output current up to 9a ? up to 95% e fficiency ? adjustable output voltage from 0.8v to 5.5v ? 1% fb accuracy ? any capacitor ? s table ? zero - to - h i gh esr ? 600khz s witching f requency ? power g ood (pg) output ? fold back current - limit and ?h iccup ? mode s hort - circuit protection ? safe start - up into pre - biased loads ? ? 40 c to +125 c junction temperature range ? available in 28 - pin 5mm 6mm qfn package application s ? servers and work stations ? routers, switches, and t elecom equipment ? base station s ___________________________________________________________________________________________________________ typical application efficiency (vin = 12v) vs. output current 50 55 60 65 70 75 80 85 90 95 100 0 2 4 6 8 10 12 output current (a) efficiency (%) 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v vin = 12v
micrel, inc. mic24054 october 2012 2 m9999 - 10 2 5 12 - a ordering information part number swit ching frequency voltage package junction temperature range lead finish mic24054 yjl 600khz adjustable 28- pin 5mm 6mm qfn ? 40 c to +125c pb - free pin configuration 28- pin 5mm x 6mm qfn (j l) ( top view ) pin description pin number pin name pin functi on 1 pvdd 5v internal linear regulator o utput . pvdd supply is the power mosfet gate drive supply voltage and created by internal ldo from v in . when v in < +5.5v , pvdd s hould be tied to pvin pins. a 2.2 f ceramic capacitor from the pvdd pin to p gnd (p in 2) must be place d next to the ic. 2, 5, 6, 7, 8, 21 pgnd power ground. pgnd is the ground path for the mic24054 buck converter power stage. the pgnd pins connect to the low - side n - channel internal mosfet gate drive supply ground, the sources of the mosfets, the negative terminals of input capacitors, and the negative terminals of output capacitors. the loop for the power ground should be as small as possible and separate from the signal ground (sgnd) loop. 3 nc no connect. 4, 9, 10, 11, 12 sw switch node o utput. internal connection for the high - side mosfet source and low - side mosfet drain. due to the high speed switching on this pin, the sw pin should be routed away from sensitive nodes. 13,14,15, 16,17,18,19 pvin high - side n - internal mosfet drain connec tion i nput . the pv in operating voltage range is from 4.5v to 19v. input capacitors between the pv in pins and the power ground (pgnd) are required and keep the connection short. 20 bst boost o utput. bootstrapped voltage to the high - side n - channel mosfet driver. a schottky diode is connected between the pvdd pin and the bst pin. a boost capacito r of 0.1f is connected between the bst pin and the sw pin. adding a small resistor at the bst pin can slow down the turn - on time of high- side n - channel mosfets.
micrel, inc. mic24054 october 2012 3 m9999 - 10 2 5 12 - a pin description (continued) pin number pin name pin function 22 cs current sense i nput . t he cs pin senses current by monitoring the voltage across the low - side mosfet during the off - time. the current sensing is necessary for short circuit protection and zero current cross comparator. in order to sense the current accurately, connect the low - side mosfet drain to sw using a kelvin connection. the cs pin is also the high - side mosfet?s output driver return. 23 sgnd signal ground . sgnd must be connected directly to the ground planes. do not route the sgnd pin to the pgnd p ad on the top layer , see pcb layout guidelines for details. 24 fb feedback i nput . input to the transconductance amplifier of the control loop. the fb pin is regulated to 0.8v. a resistor divider connecting the feedback to the output is used to adjust the desired output vol tage. 25 pg power good o utput . open drain output . the pg pin is externally tied with a resistor to vdd. a high output is asserted when v out > 92% of nominal. 26 en enable i nput . a logic level control of the output. the en pin is cmos - compatible. lo gic high = enable, logic low = shutdown. in the off state, supply current of the device is greatly reduced (typically 5a). the en pin should not be left floating . 27 vin power supply voltage i nput . requires bypass capacitor to sgnd. 28 vdd 5v interna l linear regulator o utput . vdd supply is the supply bus for the ic control circuit . vdd is created by internal ldo from vin . when v in < + 5.5v , vdd s hould be tied to pvin pins. a 1 f ceramic capacitor from the vdd pin to s gnd pins must be place next to the ic.
micrel, inc. mic24054 october 2012 4 m9999 - 10 2 5 12 - a absolute maximum ratings (1 ) pvin to pgnd ............................................... ? 0.3v to +29v vin to pgnd ................................................. ? 0.3v to pvin pvdd, vdd to pgnd ..................................... ? 0.3v to +6v v sw , v cs to pgnd ............................. ? 0.3v to (pvin +0.3v) v bst to v sw ........................................................ ? 0.3v to 6v v bst to pgnd .................................................. ? 0.3v to 35v v fb , v pg to pgnd ............................. ? 0.3v to (vdd + 0.3v) v en to pgnd ....................................... ? 0.3v to (vin +0.3v) pgnd to sgnd ............................................ ? 0.3v to +0.3v junction temperature .............................................. +150c storage temperature (t s ) ......................... ? 65 c to +150 c lead temperature (soldering, 10s) ............................ 260c esd rating (2). ................................................ esd sensitive operating ratings (3) supply voltage (pvin, vin) .............................. 4.5v to 19 v pvdd, vdd supply voltage (pvdd, vdd) ..... 4.5v to 5.5v enable input (v en ) .................................................. 0v to v in junction temperature (t j ) ........................ ? 40 c to +125 c maximum power dissipation ...................................... note 4 package thermal resistance (4) 5mm x 6mm qfn - 28 ( ja ) ................................ 28 c/w electrical characteristics (5 ) p vin = vin = v en = 12v, v bst ? v sw = 5v; t a = 25c, unless noted. bold values indicate ? 4 0c t j + 12 5c. parameter condition min . typ . max . units power supply input input voltage range (v in , pv in ) 4.5 19 v quiescent supply current v fb = 1.5v (non - switching) 450 750 a shutdown supply current v en = 0v 5 10 a vdd supply voltage vdd output voltage vin = 7v to 19v , i dd = 25 ma 4. 8 5 5.4 v vdd uvlo threshold vdd rising 3.7 4.2 4.5 v vdd uvlo hysteresis 400 mv dropout voltage (v in ? vdd) i dd = 25ma 380 600 mv dc/ dc controller output - voltage adjust range (v out ) 0.8 5.5 v reference feedback reference voltage 0 c t j 85c ( 1.0 %) 0.7 92 0.8 0.8 08 v ? 40c t j 125c ( 1.5 %) 0.788 0.8 0.812 load regulation i out = 3a to 9 a (continuous mode) 0.25 % line regulation vin = 4.5v to 19v 0.25 % fb bias current v fb = 0 .8v 50 500 na notes: 1. exceeding the absolute maximum rating may damage the device. 2. devices are esd sensitive. handling precautions recommended. human body model, 1.5k ? in series with 100pf. 3. the device is not guaranteed to function outside operating range. 4. pd (max) = (t j (max) ? t a )/ ja , where ja depends upon the printed circuit layout. a 5 square inch 4 layer, 0.62?, fr - 4 pcb with 2oz finish copper weight per layer is used for the ja . 5. specification for packaged product only .
micrel, inc. mic24054 october 2012 5 m9999 - 10 2 5 12 - a electrical characteristics (5) (continued) pvin = vin = v en = 12v, v bst ? v sw = 5v; t a = 25c, unless noted. bold values indicate ? 4 0c t j +125c. parameter condition min. typ. max. units enable control en logic level high 1.8 v en logic level low 0.6 v en bias current v en = 12v 6 30 a oscillator switching frequency (6 ) v out = 2.5v 450 600 750 khz maximum duty cycle ( 7) v fb = 0v 82 % minimum duty cycle v fb = 1.0v 0 % minimum off - time 300 n s soft - start soft - start time 3 ms short - circuit protection peak inductor current - limit threshold v fb = 0.8v, t j = 25 c 12.5 14 20 a v fb = 0.8v, t j = 125c 11.25 short - circuit current v fb = 0v 8 a internal fets top - mosfet r ds (on) i sw = 3 a 27 m ? bottom - mosfet r ds (on) i sw = 3 a 10.5 m ? sw leakage current v en = 0v 60 a v in leakage current v en = 0v 25 a power good (pg) pg threshold voltage sweep v fb from low to high 85 92 95 %v out pg hysteresis sweep v fb from high to low 5.5 %v out pg delay time sweep v fb from low to high 100 s pg low voltage sweep v fb < 0.9 v nom , i pg = 1ma 70 200 mv thermal protection over - temperature shutdown t j rising 160 c over - temperature shutdown hysteresis 15 c notes: 6. measured in t est mode. 7. the maximum duty - cycle is limited by the fixed mandatory off - time t off of typically 300ns.
micrel, inc. mic24054 october 2012 6 m9999 - 10 2 5 12 - a typical characteristics vin operating supply current vs. input voltage 0.0 0.2 0.4 0.6 0.8 1.0 4 7 10 13 16 19 input voltage (v) supply current (ma) v out = 1.8v i out = 0a switching vin shutdown current vs. input voltage 0 10 20 30 40 4 7 10 13 16 19 input voltage (v) shutdown current (a) v en = 0v r en = open vdd output voltage vs. input voltage 0 2 4 6 8 10 4 7 10 13 16 19 input voltage (v) vdd voltage (v) v fb = 0.9v i dd = 10ma feedback voltage vs. input voltage 0.792 0.796 0.800 0.804 0.808 4 7 10 13 16 19 input voltage (v) feedback voltage (v) v out = 1.8v i out = 2a total regulation vs. input voltage -1.0% -0.5% 0.0% 0.5% 1.0% 4 7 10 13 16 19 input voltage (v) total regulation (%) v out = 1.8v i out = 2a to 9a output current limit vs. input voltage 0 5 10 15 20 4 7 10 13 16 19 input voltage (v) current limit (a) v out = 1.8v switching frequency vs. input voltage 500 550 600 650 700 4 7 10 13 16 19 input voltage (v) frequency (khz) v out = 1.8v i out = 2a enable input current vs. input voltage 0 4 8 12 16 4 7 10 13 16 19 input voltage (v) en input current (a) v en = vin pg/v ref ratio vs. input voltage 80% 85% 90% 95% 100% 4 7 10 13 16 19 input voltage (v) v pg threshold/v ref (%) v fb = 0.8v
micrel, inc. mic24054 october 2012 7 m9999 - 10 2 5 12 - a typical characteristics (continued) vin operating supply current vs. temperature 0.0 0.2 0.4 0.6 0.8 1.0 -50 -25 0 25 50 75 100 125 temperature (c) supply current (ma) vin = 12v v out = 1.8v i out = 0a switching feedback voltage vs. temperature 0.788 0.792 0.796 0.800 0.804 0.808 -50 -25 0 25 50 75 100 125 temperature (c) feeback voltage (v) vin = 12v v out = 1.8v i out = 2a load regulation vs. temperature -1.0% -0.5% 0.0% 0.5% 1.0% -50 -25 0 25 50 75 100 125 temperature (c) load regulation (%) vin = 12v v out = 1.8v i out =2a to 9a line regulation vs. temperature -0.6% -0.5% -0.4% -0.3% -0.2% -0.1% 0.0% 0.1% 0.2% 0.3% -50 -25 0 25 50 75 100 125 temperature (c) line regulation (%) vin = 4.5v to 19v v out = 1.8v i out = 2a switching frequency vs. temperature 500 550 600 650 700 -50 -25 0 25 50 75 100 125 temperature (c) frequency (khz) vin = 12v v out = 1.8v i out = 2a vdd vs. temperature 2 3 4 5 6 -50 -25 0 25 50 75 100 125 temperature (c) vdd (v) vin = 12v v out = 1.8v i out =0a output current limit vs. temperature 0 5 10 15 20 -50 -25 0 25 50 75 100 125 temperature (c) current limit (a) vin = 12v v out = 1.8v 0 2 4 6 8 10 12 14 -50 -25 0 25 50 75 100 125 shutdown current ( a) temperature ( c) vin shutdown current vs. temperature vin = 12v i out = 0a v en = 0v -0.1 0.9 1.9 2.9 3.9 4.9 -50 -25 0 25 50 75 100 125 vdd threshold (v) temperature ( c) vdd uvlo threshold vs. temperature rising falling hyst
micrel, inc. mic24054 october 2012 8 m9999 - 10 2 5 12 - a typical characteristics (continued) feedback voltage vs. output current 0.792 0.796 0.800 0.804 0.808 0 1.5 3 4.5 6 7.5 9 output current (a) feedback voltage (v) vin = 12v v out = 1.8v output voltage vs. output current 1.782 1.787 1.791 1.796 1.800 1.805 1.810 1.814 1.819 0 1.5 3 4.5 6 7.5 9 output current (a) output voltage (v) vin = 12v v out = 1.8v line regulation vs. output current -1.0% -0.5% 0.0% 0.5% 1.0% 0 1.5 3 4.5 6 7.5 9 output current (a) line regulation (%) vin = 4.5v to 19v v out = 1.8v switching frequency vs. output current 500 550 600 650 700 2 4 6 8 10 output current (a) frequency (khz) vin = 12v v out = 1.8v output voltage (vin = 5v) vs. output current 3.0 3.4 3.8 4.2 4.6 5.0 0 2 4 6 8 10 12 output current (a) output voltage (v) t a 25oc 85oc 125oc vin = 5v v fb < 0.8v die temperature* (vin = 5v) vs. output current 0 20 40 60 80 0 1.5 3 4.5 6 7.5 9 output current (a) die temperature (c) vin = 5v v out = 1.8v die temperature* (vin = 12v) vs. output current 0 20 40 60 80 0 2 3 5 6 8 9 output current (a) die temperature (c) vin = 12v v out = 1.8v ic power dissipation (vin = 5v) vs. output current 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 0 1.5 3 4.5 6 7.5 9 output current (a) power dissipation (w) vin = 5v v out = 3.3v v out = 0.8v ic power dissipation (vin = 12v) vs. output current 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 0 1.5 3 4.5 6 7.5 9 output current (a) power dissipation (w) v out = 5.0v v out = 0.8v vin = 12v die temperature* : the temperature measurement was taken at the hottest point on the mic24054 case mounted on a 5 square inch 4 layer, 0.62?, fr - 4 pcb wit h 2oz finish copper weight per layer, see thermal measurement section. actual results will depend upon the size of the pcb, a mbient temperature and proximity to other heat emitting components.
micrel, inc. mic24054 october 2012 9 m9999 - 10 2 5 12 - a typical characteristics (continued) efficiency (vin = 5v) vs. output current 50 55 60 65 70 75 80 85 90 95 100 0 2 4 6 8 10 12 output current (a) efficiency (%) vin = 5v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v efficiency (vin = 12v) vs. output current 50 55 60 65 70 75 80 85 90 95 100 0 2 4 6 8 10 12 output current (a) efficiency (%) 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v vin = 12v thermal derating* vs. ambient temperature 0 2 4 6 8 10 12 14 -50 -25 0 25 50 75 100 125 ambient temperature (c) output current (a) 1.5v 0.8v v in = 5v v out = 0.8, 1.2, 1.5v thermal derating* vs. ambient temperature 0 2 4 6 8 10 12 14 -50 -25 0 25 50 75 100 125 ambient temperature (c) output current (a) 3.3v 1.8v v in = 5v v out = 1.8, 2.5, 3.3v thermal derating* vs. ambient temperature 0 2 4 6 8 10 12 14 -50 -25 0 25 50 75 100 125 ambient temperature (c) output current (a) v in = 12v v out = 0.8, 1.2, 1.8v 1.8v 0.8v thermal derating* vs. ambient temperature 0 2 4 6 8 10 12 14 -50 -25 0 25 50 75 100 125 ambient temperature (c) output current (a) v in = 12v v out = 2.5, 3.3, 5v 5v 2.5v die temp erature* : the temperature measurement was taken at the hottest point on the mic24054 case mounted on a 5 square inch 4 layer, 0.62?, fr - 4 pcb with 2oz finish copper weight per layer, see thermal measurement section. actual results will depend upon the siz e of the pcb, ambient temperature and proximity to other heat emitting components.
micrel, inc. mic24054 october 2012 10 m9999 - 10 2 5 12 - a functional characteristics
micrel, inc. mic24054 october 2012 11 m9999 - 10 2 5 12 - a functional characteristics ( c ontinued)
micrel, inc. mic24054 october 2012 12 m9999 - 10 2 5 12 - a functional characteristics ( c ontinued)
micrel, inc. mic24054 october 2012 13 m9999 - 10 2 5 12 - a fu nctional diagram figure 1. mic24054 block diagram
micrel, inc. mic24054 october 2012 14 m9999 - 10 2 5 12 - a functional description the mic24054 is a n adaptive on - time synchronous s tep - d own dc/ dc regulator with an internal 5v linear regulator and a power good (pg) output. it is designed to operate over a wi de input voltage range from 4.5 v to 19v and provides a regulated output voltage at up to 9 a of output current . a n adaptive on - time control scheme is employed in to obtain a constant switching frequency and to simplify the control compensation. o ver - current protection is implemented without the use of an external sense resistor. t he device includes an internal soft - start function which reduces the power supply input surge current at start - up by controlling the output voltage rise time. theory of operation th e mic24054 is able to operate in either continuous mode or discontinuous mode. the operating mode is determined by the output of the zero cross comparator (zc) as shown in figure 1. continuous mode in continuous mode, t he output voltage is sensed by the mi c24054 feedback pin fb via the voltage divider r1 and r2, and compared to a 0.8v reference voltage v ref at the error comparator through a low gain transconductance (g m ) amplifier. if the feedback voltage decreases and the output of t he g m amplifier is belo w 0.8v, t he n the error comparator will trigger the control logic and generate an on - time period. the on - time period length is predetermined by the ?fixed t on estimation? circuitry: 600khz v v = t in out ed) on(estimat eq. 1 where v out is the output voltage and v in i s the power stage input voltage. at the end of the on - time period, the internal high - side driver turns o f f the high - side mosfet and the low - side driver turn s on the low - side mosfet . the off - time period length depend s up on the feedback voltage in most cas es. when the feedback voltage decreases and the output of the g m amplifier is below 0.8v, the on - time period is trigger ed and the off - time period ends. if the off - time period de termined by the feedback voltage is less than the minimum off - time t off (min) , w hich is about 3 0 0ns , the mic24054 control logic will apply the t off (min) instead. t off (min) is required to maintain enough energy in the boost capacitor (c bst ) to drive the high - side mosfet . the maximum duty cycle is obtained from the 30 0ns t off (min) : s s off(min) s max t 300ns 1 t t t d - - = = eq. 2 where t s = 1/6 00khz = 1. 66 s. it is not recommended to use mic24054 with a off - time close to t off (min) during steady - state operation . also, as v out increases, the internal ripple injection will increase and reduce the line regulation performance. therefore, the maximum output vol tage of the mic24054 should be limited to 5.5v and the maximum external ripple injection should be limited to 200mv. please refer to ?setting output voltage? subsection in application information for more details. the actual on - time and resulting switching frequency will vary with the part - to - part variation in the rise and fall time s of the internal mosfets , the output load current, and variations in the v dd voltage . also, the minimum t on results in a lower switching frequency in high v in to v out ap plicatio ns, such as 18 v to 1.0v. the minimum t on measured on the mic24054 evaluation board is about 100 ns. during load transient s , the switching frequency is changed due to the varying off - time. to illustrate the control loop operation , we will analyze both the s teady - state and load transient scenario s . figure 2 shows the mic24054 control loop timing during steady - state operation . during steady - state, the g m amplifier senses the feedback voltage ripple, which is proportional to the output voltage ripple and the i nductor current ripple, to trigger the on - time period. the on - time is predetermined by the t on estimat or . the termination of the off - time is controlled by the feedback voltage . at the valley of the feedback voltage ripple, which occurs when v fb falls below v ref , the off period ends and the next on - time period is triggered through the control logic circuitry.
micrel, inc. mic24054 october 2012 15 m9999 - 10 2 5 12 - a figure 2. mic24054 control loop timing figure 3 shows the operation of the mic24054 during a load transient . the output voltage drops due to the sudden load increas e , which cause s the v fb to be less than v ref . this will cause the error comparator to trigger an on - time period. at the end of th e on - time period, a minimum off - time t off (min) is generated to charge c bst since the feedback voltage is still below v ref . then, the next on - time period is triggered due to the low feedback voltage . therefore, the switching frequency changes during the load transient , but returns to the nominal fixed frequency once the output has stabilized at the new load c urrent level . with the varying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in mic24054 converter. figure 3. mic24054 load transient response unlike true current - mode control, the mic2 4054 uses the output voltage ripple to trigger an on - time period. the output voltage ripple is proportional to the inductor current ripple if the esr of the output capacitor is large enough . the mic24054 control loop has the advantage of eliminat ing the n eed for slope comp ensation. in order to meet the stability requirements , t he mic24054 feedback voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the g m amplifier and the error comparator. the recommended f eedback voltage ripple is 20mv ~100mv . if a low - esr output capacitor is selected, then the feedback voltage ripple may be too small to be sensed by the g m amplifier and the error comparator. also, the output voltage ripple and the feedback voltage ripple ar e not necessarily in phase with the inductor current ripple if the esr of the output capacitor is very low. in these cases , ripple injection is required to ensure proper operation . please refer to ?ripple injection? subsection in application information fo r more details about the ripple injection technique . discontinuous mode in continuous mode , the inductor current is always greater than zero; however, at light loads the mic24054 is able to force the inductor current to operate in discontinuous mode. dis continuous mode is where the inductor current falls to zero , as indicated by trace (i l ) shown in figure 4. during this period , the efficiency is optimized by shutting down all the non - essential circuits and minimizing the supply current. the mic24054 wakes up and turns on the high - side mosfet when the feedback voltage v fb drop s below 0.8v. the mic24054 has a zero c ross ing c omparator that monitors the inductor current by sensing the voltage drop across the low - side mosfet during its on - time. if the v fb > 0.8 v and the inductor current goes slightly negative, then the mic24054 automatically powers down most of the ic circuitry and goes into a low - power mode. once the mic24054 goes into discontinuous mode, both lsd and hsd are low, which turns off the high - side and low - side mosfets. the load current is supplied by the output capacitors and v out drops. if the drop of v out causes v fb to go below v ref , then all the circuits will wake up into normal continuous mode. first , the b ias currents of most circuits reduced during the discontinuous mode are restored, then a t on pulse is triggered before the drivers are turned on to avoid any possible glitches. finally, the high - side driver is turned on. figure 4 shows the control loop timing in discontinuous mode.
micrel, inc. mic24054 october 2012 16 m9999 - 10 2 5 12 - a figur e 4 . mic24054 control loop timing ( disc ontinuous mode) during discontinuous mode, the zero crossing comparator and the current limit comparator are turned off. the bias current of most circuits are reduced. as a result, the total power supply current du ring discontinuous mode is only about 450 a, allowing the mic24054 to achieve high efficiency in light load applications. v dd regulator the mic24054 provides a 5v regulated output for input voltage v in ranging from 5.5v to 19v . when v in < 5.5v, v dd should be tied to pv in pins to bypass the internal linear regulator . soft - start soft - start reduces the power supply input surge current at startup by controlling the output voltage rise time. the input surge appears while the output capacitor is charged up. a sl ower output rise time will draw a lower input surge current. the mic24054 implements an internal digital soft - start by making the 0.8v reference voltage v ref ramp fr om 0 to 100% in about 3 ms with 9.7mv step s . therefore, the output voltage is controlled to increase slowly by a stair - case v fb ramp. once the soft - start cycl e ends, the related circuitry is disabled to reduce current consumption. v dd must be powered up at the same time or after v in to make the soft - start function correctly. current limit the m ic24054 uses the r ds(on) of the internal low - side power mosfet to sense over - current conditions. this method will avoid adding cost, board space and power losses taken by a discrete current sense resistor . the low - side mosfet is used because it displays mu ch lower parasitic oscillations during switching than the high - side mosfet. in each switching cycle of the mic24054 converter, the inductor current is sensed by monitoring the low - side mosfet in the off period. if the inductor current is greater than 14a , then the mic24054 turns off the high - side mosfet and a soft - start sequence is trigge re d. this mode of operation is called ?hiccup mode? and its purpose is to protect the down stream load in case of a hard short. the load current - limit threshold has a fold back characteristic related to the feedback voltage a s shown in f igure 5 . current limit threshold vs. feedback voltage 0 4 8 12 16 20 0.0 0.2 0.4 0.6 0.8 1.0 feedback voltage (v) current limit threshold (a) figure 5. mic24054 current - limit foldback characteristic power - good (pg) the power good (pg) pin is an open drain output which indicates logic high when the output is nominally 92% of its steady state voltage. a pull - up resistor of more than 10k should be connected from pg to vdd.
micrel, inc. mic24054 october 2012 17 m9999 - 10 2 5 12 - a mosfet gate drive the block diagram ( figure 1 ) shows a bootstrap circuit, consisting of d1 (a schottky diode is recommended) and c bst . this circuit supplies energy to the high - side drive circuit. capacitor c bst is ch arged , while the low - side mosfet is on , and the voltage on the sw pin is approximately 0v. when the high - side mosfet driver is turned on, energy from c bst is used to turn the mosfet on. as the high - side mosfet turns on, the voltage on the sw pin increases to approximately v in . diode d1 is reverse biased and c bst floats high while continuing to keep the high - side mosfet on. the bias current of the high - side driver is less than 10ma so a 0.1f to 1f is sufficient to hold the gate voltage with minimal droop for the power stroke (high - side switching ) cycle, i.e. bs t = 10ma x 1.67s/0.1f = 167 mv. when the low - side mosfet is turned back on, c bst is recharged through d1. a small resistor r g , which is in series with c bst , can be used to slow down the turn - on time of the high - side n - channel mosfet. the drive voltage is derived from the v dd supply voltage . the nominal low - side gate drive voltage is v dd and the nominal high - side gate drive voltage is approximately v dd ? v diode , where v diode is the voltage drop across d1. an approximate 30ns delay between the high - side and low - side driver transitions is used to prevent current from simultaneously flowing unimpeded through both mosfets.
micrel, inc. mic24054 october 2012 18 m9999 - 10 2 5 12 - a application information inductor selection values for inductance, peak, and rms currents are required to select th e output inductor. the input and output voltages and the inductance value determine the peak - to - peak inductor ripple current. generally, higher inductance values are used with higher input voltages. larger peak - to - peak ripple currents will increase the pow er dissipation in the inductor and mosfets. larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. smaller peak - to - peak ripple currents require a larger inductance value and therefore a larger and m ore expensive inductor. a good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. the inductance value is c alculated by equation 3 : out(max) sw in(max) out in(max) out i 20% f v ) v (v v l ? = eq. 3 where: f sw = s witching frequency, 6 00khz 20% = ratio of ac ripple current to dc output current v in (max) = maximum power stage input voltage the peak - to - peak inductor current ripple is: l f v ) v (v v i sw in(max) out in(max) out l(pp) ? = ? eq. 4 the peak inductor current is equal to the average ou tput current plus one half of the peak - to - peak inductor current ripple. i l(pk) =i out(max) + 0.5 i l(pp) eq. 5 the rms inductor current is used to calculate the i 2 r losses in the inductor. 12 i i i 2 l(pp) 2 out(max) l(rms) + = eq. 6 maximiz ing efficiency requires the proper selection of core material and minimizing the winding resistance. the high frequency operation of the mic24054 requires the use of ferrite materials for all but the most cost sensitive applications. lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. this is especially noticeable at low output power. the winding resistance decreases efficiency at the higher output current levels. the winding resistance must be minimized although this usually comes at the expense of a larger inductor. the power dissipated in the inductor is equal to the sum of the core and copper losses. at higher output loads, the core losses are usually insignificant and can be ignored. at lower output currents, the core losses can be a significant contributor. core loss information is usually available from the magnetics vendor. copper loss in the inductor is calculated by equation 7 : p inductor (cu) = i l(rms) 2 r winding eq. 7 the resista nce of the copper wire, r winding , increases with the temperature. the value of the winding resistance used should be at the operating temperature. p winding (ht) = r winding (20 c) (1 + 0.0042 (t h ? t 20 c )) eq. 8 where: t h = temperature of wire under fu ll load t 20c = ambient temperature r winding(20c) = room temperature winding resistance (usually specified by the manufacturer) output capacitor selection the type of the output capacitor is usually determined by its equivalent series resistance ( e sr ). v oltage and rms current capability are two other important factors for selecting the output capacitor. recommended ca pacitor types are tantalum, low - esr aluminum electrolytic, os - con and poscap . the output capacitor?s esr is usually the main cause of the ou tput ripple. the output capacitor esr also affects the control loop from a stability point of view.
micrel, inc. mic24054 october 2012 19 m9999 - 10 2 5 12 - a the maximum value of esr is calculated: l(pp) out(pp) c i v esr out eq. 9 where: ? v out(pp) = peak - to - peak output voltage ripple i l(pp) = peak - to - peak inductor current ripple the total output ripple is a combination of the esr and output capacitance. the total ripple is calculated in equation 10 : ( ) 2 c l(pp) 2 sw out l(pp) out(pp) out esr i 8 f c i v + ? ? ? ? ? ? ? ? = eq. 10 w here : d = d uty cycle c out = o utput capacitance value f sw = s witching frequency as described in the ?theory of operation? subsection in the functional description section , the mic24054 requires at least 20mv peak - to - peak ripple at the fb pin to make the g m amp lifier and the error comparator behav e properly. also, the output voltage ripple should be in phase with the inductor current. therefore , the output voltage ripple caused by the output capacitor s value should be much smaller than the ripple caused by the o utput capacitor esr. i f low - esr capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection method should be applied to provi de the enough feedback voltage ripple . please refer to the ?ripple injection? subsection for more details. the voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or os - con. the output capacitor rms current is calculated below: 12 i i l(pp) (rms) c out = eq. 11 the power dissi pated in the output capacitor is: out out out c 2 (rms) c ) diss(c esr i p = eq. 12 input capacitor selection the input capacitor for the power stage input v in should be selected for ripple current rating and voltage rating. tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. a tantalum input capacitor?s voltage rating should be at least two times the maximum input voltage to maximize reliability. aluminum electrolytic, os - con, and multilayer polymer film capacito rs can handle the higher inrush currents without voltage de - rating. the input voltage ripple will primarily depend on the input capacitor?s esr. the peak input current is equal to the peak inductor current, so: v in = i l(pk) esr cin eq. 13 the input ca pacitor must be rated for the input current ripple. the rms value of input capacitor current is determined at the maximum output current. assuming the peak - to - peak inductor current ripple is low: d) (1 d i i out(max) cin(rms) ? eq. 14 the power dissipated in th e input capacitor is: p diss(cin) = i cin(rms) 2 esr cin eq. 15 ripple injection the v fb ripple re quired for proper operation of the mic24054 g m amplifier and error comparator is 20mv to 100mv . however, the output voltage ripple is generally designed as 1 % to 2% of the output voltage. for a low output voltage, such as a 1v, the output voltage ripple is only 10mv to 20mv, and the feedback voltage ripple is less than 20mv. if the feedback voltage ripple is so small that the g m amplifier and error comparator can ?t sense it, then the mic24054 will lose control and the output voltage is not regulated. in order to have some amount of v fb ripple, a ripple injection method is applied for low output voltage ripple applications.
micrel, inc. mic24054 october 2012 20 m9999 - 10 2 5 12 - a the applications are divided into thr ee situations according to the amount of the feedback voltage ripple: 1. enough ripple at the feedback voltage due to the large esr of the output capacitors. as shown in figure 6 , the converter is stable without any ripple injection . the feedback voltage rip ple is: (pp) l c fb(pp) i esr r2 r1 r2 v out + = eq. 16 w here : i l(pp) is the peak - to - peak value of the inductor current ripple. 2. inadequate ripple at the feedback voltage due to the small esr of the output capacitors. the output voltage ripple is fed into the fb pin through a feedforward capacitor c ff in this situat ion, as shown in figure 7 . the typical c ff value is between 1nf and 100nf. with the feedforward capacitor, the feedback voltage ripple is very close to the output voltage ripple: (pp) l fb(pp) i esr v eq. 17 3. virtually no ripple at the fb pin voltage d ue to the very low esr of the output capacitors. figure 6 . enough ripple at fb figure 7 . inadequate ripple at fb figure 8 . invisible ripple at fb in this situation, the output voltage ripple is less than 20mv. therefore, additional ripple is injected into the fb pin from the switching node sw via a resistor r inj and a capacitor c inj , as shown in figure 8 . the injected ripple is: = sw div in fb(pp) f 1 d) - (1 d k v v eq. 18 r1//r2 r r1//r2 k inj div + = eq. 19 w here : v in = power stage input voltage d = d uty c ycle f sw = s witching frequency 2 = (r1//r2//r inj ) c ff in e quations 18 and 19 , it is assumed that the time constant associated with c ff must be much greater than the switching period: 1 t f 1 sw << = eq. 20 if the voltage divider resistors r1 and r2 are in the k range, a c ff of 1nf to 100nf can easily satisfy the large time constant requirements . also, a 100nf injection capacitor c inj is used in order to be considered as short for a wide range of the frequencies.
micrel, inc. mic24054 october 2012 21 m9999 - 10 2 5 12 - a the process of sizing the ripple injection resistor and cap acitors is: step 1. select c ff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. typical choice of c ff is 1nf to 100nf if r1 and r2 are in k range. step 2. select r inj according to the expected feedback v oltage ripple using equation 19 : d) (1 d f v v k sw in fb(pp) div - = eq. 21 then the value of r inj is obtained as: 1) k 1 ( (r1//r2) r div inj ? = eq. 22 step 3. select c inj as 100nf, which could be considered as short for a wide range of the frequencies. setting output voltage the mic24054 requires two resistors to set the out put voltage as shown in figure 9 . the outpu t voltage is determined by e quation 23: ) r2 r1 (1 v v fb out + = eq. 23 where: v fb = 0.8v. a typical va lue of r1 can be between 3k and 10k. if r1 is too large, it may allow noise to be introduced into the voltage feedback loop. if r1 is too small, it will decrease the efficiency of the power supply, especially at light loads. once r1 is selected, r2 can b e calculated using equation 24 : fb out fb v v r1 v r2 ? = eq. 24 figure 9 . voltage - divider configuration in addition to the external ripple injection added at the fb pin, internal ripple injection is added at the inverting input of the comparator insid e the mic24054 , as shown in figure 10 . the inverting input voltage v inj is clamped to 1.2v. as v out is increased, the swing of v inj will be clamped. the clamped v inj reduces the line regulation because it is reflected as a dc error on the fb terminal. ther efore, the maximum output voltage of the mic24054 should be limited to 5.5v to avoid this problem. figure 10 . internal ripple injection thermal measurements measuring the ic?s case temperature is recommended to insure it is within its operating limit s. although this might seem like a very elementary task, it is easy to get erroneous results. the most common mistake is to use the standard thermal couple that comes with a thermal meter. this thermal couple wire gauge is large, typically 22 gauge, and be haves like a heatsink, resulting in a lower case measurement. two methods of temperature measurement are using a smaller thermal couple wire or an infrared thermometer. if a thermal couple wire is used, then it must be constructed of 36 gauge wire or higher the n (smaller wire size) to minimize the wire heat - sinking effect. in addition, the thermal couple tip must be covered in either thermal grease or thermal glue to make sure that the thermal couple junction is making good contact with the case of the ic. omega brand thermal couple (5sc - tt - k - 36- 36) is adequate for most applications. wherever possible, an infrared thermometer is recommended. the measurement spot size of most infrared thermometers is too large for an accurate reading on a small form factor ics. h owever, a ir thermometer from optris has a 1mm spot size, which makes it a good choice for measuring the hottest point on the case. an optional stand makes it easy to hold the beam on the ic for long periods of time.
micrel, inc. mic24054 october 2012 22 m9999 - 10 2 5 12 - a pcb layout guideline s warning!!! to mi nimize emi and output noise, follow these layout recommendations. pcb layout is critical to achieve reliable, stable and efficient performance. a ground plane is required to control emi and minimize the inductance in power, signal and return paths. the fol lowing guidelines should be followed to insure proper operation of the mic24054 regulator . ic ? a 2.2f ceramic capacitor , which is connect ed to the p v dd pin , must be located right at the ic. the p v dd pin is very noise sensitive and placement of the capacito r is very critical. use wide traces to connect to the p v dd and pgnd pins. ? a 1 f ceramic capacitor must be placed right between vdd and the signal ground sgnd. the s gnd must be connected directly to the ground planes. do not route the s gnd pin to the pgnd p ad on the top layer. ? place the ic close to the point - of - load (pol). ? use fat traces to route the input and output power lines. ? signal and power grounds should be kept separate and connected at only one location. input capacitor ? place the input capacitor ne xt. ? place the input capacitors on the same side of the board and as close to the ic as possible. ? keep both the pvin p in and pgnd connections short. ? place several vias to the ground plane close to the input capacitor ground terminal. ? use either x7r or x5r d ielectric input capacitors. do not use y5v or z5u type capacitors. ? do not replace the ceramic input capacitor with any other type of capacitor. any type of capacitor can be placed in parallel with the input capacitor. ? if a tantalum input capacitor is place d in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. ? in ?hot - plug? applications, a tantalum or electrolytic bypass capacitor must be used to limit the over - vol tage spike seen on the input supply with power is suddenly applied. inductor ? keep the inductor connection to the switch node ( sw ) short. ? do not route any digital lines underneath or close to the inductor. ? keep the switch node ( sw ) away from the feedback (fb) pin. ? the cs pin should be connected directly to the sw pin to accurate sense the voltage across the low - side mosfet. ? to minimize noise, place a ground plane underneath the inductor. ? the inductor can be placed on the opposite side of the pcb with resp ect to the ic. it does not matter whether the ic or inductor is on the top or bottom as long as there is enough air flow to keep the power components within their temperature limits. the input and output capacitors must be placed on the same side of the bo ard as the ic. output capacitor ? use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. ? phase margin will change as the output capacitor value and esr changes. contact the factory if the output capacitor is different from what is shown in the bom. ? the feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. sensing a long high current load trace can degrade the dc load regulation. optional rc snubber ? place the rc snubber on either side of the board and as close to the sw pin as possible.
micrel, inc. mic24054 october 2012 23 m9999 - 10 2 5 12 - a evaluation board schematic figure 11 . schematic of mic24054 evaluation board (j11 , r13, r15 are for testing purposes)
micrel, inc. mic24054 october 2012 24 m9999 - 10 2 5 12 - a evaluation board schematic (continued) figure 12 . schematic of mic24054 evaluation board (j11, r13, r15 are for testing purposes) (optimized for smaller footprint)
micrel, inc. mic24054 october 2012 25 m9999 - 10 2 5 12 - a bill of materials item part number manufacturer description qty . c1 open c2 , c3 12103c475kat2a avx ( 1 ) 4.7f ceramic cap acitor, x 7 r, size 1210, 25 v 2 grm32dr71e475ka61k murata ( 2 ) c3225x7r1e475k tdk ( 3 ) c13, c15 open c4, c5 12106d107mat2a avx 100 f ceramic capacitor, x5r, size 1210, 6.3v 2 grm32er60j107me20l murata c3225x5r0j107m tdk c6, c7, c10 06035c 104kat2a avx 0.1f ceramic capacitor, x7r, size 0603, 50v 3 grm188r71h104ka93d murata c1608x7r1h104k tdk c8 0603z c 105 kat 2a avx 1.0f ceramic capacitor, x7r, size 0603 , 10v 1 grm 188r7 1a105k a61d murata c 1608x7 r1a105k tdk c9 0603zd225k at2a avx 2.2f ceramic capacitor, x5 r, size 0603 , 10v 1 grm188r61a225ke34d murata c1608 x 5 r1a225k tdk c12 06035c472 kaz2a avx 4.7 nf ceramic capacitor, x7r, size 0603, 50v 1 grm188r71h472 k murata c1608x7r1h472 k tdk c14 b41851f7227m epcos ( 4 ) 220 f aluminum capacitor, 35 v 1 c11, c16 open d1 sd103 a ws mcc ( 5 ) 40v, 350ma, schottky diode, sod323 1 sd103 a ws -7 d iodes i nc ( 6 ) sd103 a ws vishay ( 7 ) l1 hcf1305 - 2r2 - r cooper bussmann ( 8 ) 2.2 h inductor, 15 a saturation current 1 r1 crcw06032r21fkea vish ay dale 2.21 ? resistor, size 0603, 1% 1 r2 crcw0 6032 r 00 fkea vishay dale 2.00 ? resistor , size 0 603 , 1% 1 r3 crcw060319k6 fkea vishay dale 19.6 k ? resistor, size 0603, 1% 1 r4 crcw06032k49 fkea vishay dale 2.49 k ? resistor, size 0603, 1% 1 r 5 crcw0603 20k0 fke a vishay dale 20.0 k ? resistor, size 0603, 1% 1 r 6, r14, r17 crcw 060310k0 fkea vishay dale 10.0k ? resistor, size 0603, 1% 3 r 7 crcw0603 4k99 fkea vishay dale 4.99 k ? resistor, size 0603, 1% 1 r 8 crcw06032k87 fkea vishay dale 2.87 k ? resistor, size 0603, 1% 1 r 9 crcw 06032k006 fkea vishay dale 2.00k ? resistor, size 0603, 1% 1 r 10 crcw0603 1k18 fkea vishay dale 1.18 k ? resistor, size 0603, 1% 1 r 11 crcw0603 806r fkea vishay dale 806? resistor, size 0603, 1% 1 r 12 crcw 0603475r fkea vishay dale 475 ? resistor, size 0603 , 1% 1
micrel, inc. mic24054 october 2012 26 m9999 - 10 2 5 12 - a bill of materials (continued) item part number manufacturer description qty r 13 crcw06030000 fk ea vishay dale 0 ? resistor, size 0603, 5 % 1 r 15 crcw0603 49r9 fkea vishay dale 49.9 ? resistor, size 0603, 1% 1 r16, r18 crcw0 603 1r21fkea vishay dale 1.21 ? resistor , size 0 603 , 1% 2 r20 open all reference designators ending with ?a? open u1 mic24054 y jl micrel. inc. ( 9 ) 12 v , 9a high - e fficiency buck regulator 1 notes: 1. avx: www.avx.com . 2. murata: www.murata.com . 3. tdk: www.tdk.com . 4. epcos: www.epcos.com . 5. mcc : www.mccsemi.com . 6. diode inc.: www.diodes.com . 7. vishay: www.vishay.com . 8. cooper bussmann : www.cooperbussmann.com . 9. micrel, inc.: www.micrel.com .
micrel, inc. mic24054 october 2012 27 m9999 - 10 2 5 12 - a recommended pcb layout figure 13 . mic24054 evaluation board top layer figure 14 . mic24054 evaluation board mid - layer 1 (ground plane)
micrel, inc. mic24054 october 2012 28 m9999 - 10 2 5 12 - a recommended pcb layout (continued) figure 1 5 . mic24054 evaluation board mid - layer 2 figure 1 6 . m ic24054 evaluation board bottom layer
micrel, inc. mic24054 october 2012 29 m9999 - 10 2 5 12 - a package information (1) 28- pin 5mm 6mm qfn (jl) note: 1. package information is correct as of the publication date. for updates and most current information, go to www.micrel.com .
micrel, inc. mic24054 october 2012 30 m9999 - 10 2 5 12 - a micrel, inc. 2180 fortune drive san jose, ca 95131 usa tel +1 (408) 944 - 0800 fax +1 (408) 474 - 1000 web http://www.micrel.com micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. this information is not intended as a warranty and micrel does not assume responsibility for its use. micrel reserves the right to change circuitry, specifications and descriptions at any time without notice. no license, whether express, implied, arising by estoppel or otherwise, to any intellectual property rights is granted by this document. except as provided in micrel?s terms and conditio ns of sale for such products, micrel assumes no liability whatsoever, and micrel disclaims any express or implied warranty relating to the sale and/or use of micrel products including liability or warranties relating to fitness for a particular purpose, me rchantability, or infringement of any patent, copyright or other intellectual property right . micrel products are not designed or authorized for use as components in life support appliances, devices or systems where mal function of a product can reasonably be expected to result in personal injury. life support devices or systems are devices or systems that (a) are intended for su rgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. a purchaser?s use or sale of micrel products for use in life support appliances, devices or systems is a purchaser?s own risk a nd purchaser agrees to fully indemnify micrel for any damages resulting from such use or sale. ? 2012 micrel, incorporated.

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